Sigma Delta Over-Sampling Charge Pump Analog-To-Digital Converter

ABSTRACT

Techniques are presented for determining current levels based on the behavior of a charge pump system while driving a load under regulation. While driving the load under regulation, the number of pump clocks during a set interval is counted. This can be compared to a reference that can be obtained, for example, from the numbers of cycles needed to drive a known load current over an interval of the duration. By comparing the counts, the amount of current being drawn by the load can be determined. This technique can be applied to determining leakage from circuit elements, such as word lines in a non-volatile memory. The accuracy and level of resolution can be further increased through use of sigma delta noise shaping.

CROSS-REFERENCE TO RELATED APPLICATION

This application is a Continuation-in-Part of U.S. patent application Ser. No. 13/628,465, filed on Sep. 27, 2012, which is hereby fully incorporated into the present application by this reference.

FIELD OF THE INVENTION

This invention relates generally to the determining current levels and, more specifically, to the use of the regulation behavior of charge pumps for determining current levels.

BACKGROUND OF THE INVENTION

Solid-state memory capable of nonvolatile storage of charge, particularly in the form of EEPROM and flash EEPROM packaged as a small form factor card, has recently become the storage of choice in a variety of mobile and handheld devices, notably information appliances and consumer electronics products. Unlike RAM (random access memory) that is also solid-state memory, flash memory is non-volatile and retains its stored data even after power is turned off. In spite of the higher cost, flash memory is increasingly being used in mass storage applications. Conventional mass storage, based on rotating magnetic medium such as hard drives and floppy disks, is unsuitable for the mobile and handheld environment. This is because disk drives tend to be bulky, are prone to mechanical failure and have high latency and high power requirements. These undesirable attributes make disk-based storage impractical in most mobile and portable applications. On the other hand, flash memory, both embedded and in the form of a removable card, are ideally suited in the mobile and handheld environment because of its small size, low power consumption, high speed and high reliability features.

EEPROM and electrically programmable read-only memory (EPROM) are non-volatile memory that can be erased and have new data written or “programmed” into their memory cells. Both utilize a floating (unconnected) conductive gate, in a field effect transistor structure, positioned over a channel region in a semiconductor substrate, between source and drain regions. A control gate is then provided over the floating gate. The threshold voltage characteristic of the transistor is controlled by the amount of charge that is retained on the floating gate. That is, for a given level of charge on the floating gate, there is a corresponding voltage (threshold) that must be applied to the control gate before the transistor is turned “on” to permit conduction between its source and drain regions.

The floating gate can hold a range of charges and therefore can be programmed to any threshold voltage level within a threshold voltage window. The size of the threshold voltage window is delimited by the minimum and maximum threshold levels of the device, which in turn correspond to the range of the charges that can be programmed onto the floating gate. The threshold window generally depends on the memory device's characteristics, operating conditions and history. Each distinct, resolvable threshold voltage level range within the window may, in principle, be used to designate a definite memory state of the cell. When the threshold voltage is partitioned into two distinct regions, each memory cell will be able to store one bit of data. Similarly, when the threshold voltage window is partitioned into more than two distinct regions, each memory cell will be able to store more than one bit of data.

In the usual two-state EEPROM cell, at least one current breakpoint level is established so as to partition the conduction window into two regions. When a cell is read by applying predetermined, fixed voltages, its source/drain current is resolved into a memory state by comparing with the breakpoint level (or reference current IREF). If the current read is higher than that of the breakpoint level, the cell is determined to be in one logical state (e.g., a “zero” state). On the other hand, if the current is less than that of the breakpoint level, the cell is determined to be in the other logical state (e.g., a “one” state). Thus, such a two-state cell stores one bit of digital information. A reference current source, which may be externally programmable, is often provided as part of a memory system to generate the breakpoint level current.

In order to increase memory capacity, flash EEPROM devices are being fabricated with higher and higher density as the state of the semiconductor technology advances. Another method for increasing storage capacity is to have each memory cell store more than two states.

For a multi-state or multi-level EEPROM memory cell, the conduction window is partitioned into more than two regions by more than one breakpoint such that each cell is capable of storing more than one bit of data. The information that a given EEPROM array can store is thus increased with the number of states that each cell can store, EEPROM or flash EEPROM with multi-state or multi-level memory cells have been described in U.S. Pat. No. 5,172,338.

The transistor serving as a memory cell is typically programmed to a “programmed” state by one of two mechanisms. In “hot electron injection,” a high voltage applied to the drain accelerates electrons across the substrate channel region. At the same time a high voltage applied to the control gate pulls the hot electrons through a thin gate dielectric onto the floating gate. In “tunneling injection,” a high voltage is applied to the control gate relative to the substrate. In this way, electrons are pulled from the substrate to the intervening floating gate.

The memory device may be erased by a number of mechanisms. For EPROM, the memory is bulk erasable by removing the charge from the floating gate by ultraviolet radiation. For EEPROM, a memory cell is electrically erasable, by applying a high voltage to the substrate relative to the control gate so as to induce electrons in the floating gate to tunnel through a thin oxide to the substrate channel region (i.e., Fowler-Nordheim tunneling.) Typically, the EEPROM is erasable byte by byte. For flash EEPROM, the memory is electrically erasable either all at once or one or more blocks at a time, where a block may consist of 512 bytes or more of memory.

The memory devices typically comprise one or more memory chips that may be mounted on a card. Each memory chip comprises an array of memory cells supported by peripheral circuits such as decoders and erase, write and read circuits. The more sophisticated memory devices operate with an external memory controller that performs intelligent and higher level memory operations and interfacing.

There are many commercially successful non-volatile solid-state memory devices being used today. These memory devices may be flash EEPROM or may employ other types of nonvolatile memory cells. Examples of flash memory and systems and methods of manufacturing them are given in U.S. Pat. Nos. 5,070,032, 5,095,344, 5,315,541, 5,343,063, and 5,661,053, 5,313,421 and 6,222,762. In particular, flash memory devices with NAND string structures are described in U.S. Pat. Nos. 5,570,315, 5,903,495, 6,046,935. Also nonvolatile memory devices are also manufactured from memory cells with a dielectric layer for storing charge. Instead of the conductive floating gate elements described earlier, a dielectric layer is used. Such memory devices utilizing dielectric storage element have been described by Eitan et al., “NROM: A Novel Localized Trapping, 2-Bit Nonvolatile Memory Cell,” IEEE Electron Device Letters, vol. 21, no. 11, November 2000, pp. 543-545. An ONO dielectric layer extends across the channel between source and drain diffusions. The charge for one data bit is localized in the dielectric layer adjacent to the drain, and the charge for the other data bit is localized in the dielectric layer adjacent to the source. For example, U.S. Pat. Nos. 5,768,192 and 6,011,725 disclose a nonvolatile memory cell having a trapping dielectric sandwiched between two silicon dioxide layers. Multi-state data storage is implemented by separately reading the binary states of the spatially separated charge storage regions within the dielectric.

Defects often occur in such memory systems, both as part of the manufacturing process as well over the operating life of the device. One of the sources of such defects are the word-lines of such memory arrays, due both to word-line leakage (to another work-line or to the substrate) and to broken word-lines. These word-line related problems typically become more and more acute as device sizes scale down. Some word-line to word-line leakage does not manifest itself when the device is fresh, but only results in a failure after the stress of a number of program-erase cycles. This leakage will cause the faulty word-line to fail to program and corresponding data will be corrupted. A broken word-line will have a high resistive connection, as a result of which the cells on far end of the break will see a voltage drop during program and verify operations. As a result, the threshold voltage distribution for the broken word-line will show un-distinguishable states. Consequently, both of these sorts of defects can be detrimental to memory operation if not detected.

SUMMARY OF INVENTION

A first set of aspects relate to a circuit connectable to a load to determine the amount of current drawn by the load. The circuit includes a charge pump system, an output noise shaping circuit, and an analog to digital converter detection circuit. The charge pump system includes a charge pump connected to receive a clock signal and having an output connectable to provide an output current to the load and regulation circuitry connected to receive feedback from the output of the charge pump and regulate the charge pump by varying the clock signal based upon the feedback. The output noise shaping circuit is connected between the feedback path of the regulation circuitry and the output of the charge pump to provide current feedback to the output of the charge pump to remove low frequency noise. The analog to digital converter detection circuit is connectable to the charge pump system to determine the number of cycles of the clock signal supplied to the charge pump during an interval of a first duration while the charge pump is under regulation and, from the determined number of cycles, determine an amount of current being drawn by the load.

Various aspects, advantages, features and embodiments of the present invention are included in the following description of exemplary examples thereof, whose description should be taken in conjunction with the accompanying drawings. All patents, patent applications, articles, other publications, documents and things referenced herein are hereby incorporated herein by this reference in their entirety for all purposes. To the extent of any inconsistency or conflict in the definition or use of terms between any of the incorporated publications, documents or things and the present application, those of the present application shall prevail.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 illustrates schematically the functional blocks of a non-volatile memory chip in which the present invention may be implemented.

FIG. 2 illustrates schematically a non-volatile memory cell.

FIG. 3 illustrates the relation between the source-drain current I_(D) and the control gate voltage V_(CG) for four different charges Q1-Q4 that the floating gate may be selectively storing at any one time.

FIG. 4 illustrates an example of an NOR array of memory cells.

FIG. 5A illustrates schematically a string of memory cells organized into an NAND string.

FIG. 5B illustrates an example of an NAND array 200 of memory cells, constituted from NAND strings 50 such as that shown in FIG. 5A.

FIG. 6 illustrates the Read/Write Circuits 270A and 270B, shown in FIG. 1, containing a bank of p sense modules across an array of memory cells.

FIG. 7 illustrates schematically a preferred organization of the sense modules shown in FIG. 6.

FIG. 8 illustrates in more detail the read/write stacks shown in FIG. 7.

FIGS. 9(0)-9(2) illustrate an example of programming a population of 4-state memory cells.

FIGS. 10(0)-10(2) illustrate an example of programming a population of 8-state memory cells.

FIG. 11 illustrates a conventional technique for programming a 4-state memory cell to a target memory state.

FIG. 12 shows a circuitry detail on how voltages are supplied to word-lines.

FIG. 13 is a block diagram of an exemplary charge pump circuit.

FIG. 14 adds leakage detection circuitry to FIG. 13.

FIG. 15 illustrates the phases of the exemplary leakage detection operation.

FIG. 16 shows the current path in a calibration process for the word-line leakage process.

FIG. 17 illustrates the phases of the calibration operation.

FIG. 18 shows the distribution of memory cell threshold voltage values to illustrate symptoms of a broken word-line.

FIG. 19 illustrates the variation in the number of programming pulse-verify iterations over different word-lines.

FIG. 20 is a timing diagram for a broken word-line detection routine.

FIGS. 21A and 21B illustrate differing placements of word-line drivers.

FIGS. 22 and 23A are flows for a scan of failed bits in a program operation.

FIG. 23B is a flow for a scan of failed bits in a program operation that also includes broken word-line detection.

FIG. 24 illustrates a current based comparison for leakage determination where two different arrays are used, one unselected and for reference use and one with an erase block selected for testing.

FIG. 25 illustrates the basic operation of exemplary circuitry for determining a leakage current level.

FIG. 26 shows the elements from FIG. 25 along with some to the other elements used in the exemplary embodiment for the leakage determination circuitry.

FIG. 27 is a block diagram to schematically illustrate the relationship of the elements of FIG. 26 to two planes.

FIG. 28 is a timing diagram for one particular implementation of the leakage determination operation.

FIG. 29 adds leakage current determination elements to the part of the waveform of FIG. 28.

FIG. 30 is plot of load versus pulse count for a charge pump system driving a load under regulation within a fixed duration.

FIG. 31 schematically illustrates a charge pump drive a load.

FIG. 32 schematically illustrates a charge pump drive twice the load of FIG. 31.

FIG. 33 is a box diagram of a charge pump system with a current level detection block.

FIG. 34 is a timing diagram for an exemplary embodiment of a pump-based current determination arrangement.

FIG. 35 looks at the elements of FIG. 33 in some detail.

FIG. 36 shows a first exemplary embodiment of a charge pump system using sigma delta noise shaping.

FIG. 37 shows how the feedback current is summed with signal to remove the low frequency noise away from the consecutive samplings.

FIG. 38 shows a second exemplary embodiment of a charge pump system using sigma delta noise shaping.

FIG. 39 looks at the relationship between the regulated voltage at the output of the pump system and the pump clock.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS Memory System

FIG. 1 to FIG. 11 illustrate example memory systems in which the various aspects of the present invention may be implemented.

FIG. 1 illustrates schematically the functional blocks of a non-volatile memory chip in which the present invention may be implemented. The memory chip 100 includes a two-dimensional array of memory cells 200, control circuitry 210, and peripheral circuits such as decoders, read/write circuits and multiplexers.

The memory array 200 is addressable by word lines via row decoders 230 (split into 230A, 230B) and by bit lines via column decoders 260 (split into 260A, 260B) (see also FIGS. 4 and 5.) The read/write circuits 270 (split into 270A, 270B) allow a page of memory cells to be read or programmed in parallel. A data I/O bus 231 is coupled to the read/write circuits 270.

In a preferred embodiment, a page is constituted from a contiguous row of memory cells sharing the same word line. In another embodiment, where a row of memory cells are partitioned into multiple pages, block multiplexers 250 (split into 250A and 250B) are provided to multiplex the read/write circuits 270 to the individual pages. For example, two pages, respectively formed by odd and even columns of memory cells are multiplexed to the read/write circuits.

FIG. 1 illustrates a preferred arrangement in which access to the memory array 200 by the various peripheral circuits is implemented in a symmetric fashion, on opposite sides of the array so that the densities of access lines and circuitry on each side are reduced in half. Thus, the row decoder is split into row decoders 230A and 230B and the column decoder into column decoders 260A and 260B. In the embodiment where a row of memory cells are partitioned into multiple pages, the page multiplexer 250 is split into page multiplexers 250A and 250B. Similarly, the read/write circuits 270 are split into read/write circuits 270A connecting to bit lines from the bottom and read/write circuits 270B connecting to bit lines from the top of the array 200. In this way, the density of the read/write modules, and therefore that of the sense modules 380, is essentially reduced by one half.

The control circuitry 110 is an on-chip controller that cooperates with the read/write circuits 270 to perform memory operations on the memory array 200. The control circuitry 110 typically includes a state machine 112 and other circuits such as an on-chip address decoder and a power control module (not shown explicitly). The state machine 112 provides chip level control of memory operations. The control circuitry is in communication with a host via an external memory controller.

The memory array 200 is typically organized as a two-dimensional array of memory cells arranged in rows and columns and addressable by word lines and bit lines. The array can be formed according to an NOR type or an NAND type architecture.

FIG. 2 illustrates schematically a non-volatile memory cell. The memory cell 10 can be implemented by a field-effect transistor having a charge storage unit 20, such as a floating gate or a dielectric layer. The memory cell 10 also includes a source 14, a drain 16, and a control gate 30.

There are many commercially successful non-volatile solid-state memory devices being used today. These memory devices may employ different types of memory cells, each type having one or more charge storage element.

Typical non-volatile memory cells include EEPROM and flash EEPROM. Examples of EEPROM cells and methods of manufacturing them are given in U.S. Pat. No. 5,595,924. Examples of flash EEPROM cells, their uses in memory systems and methods of manufacturing them are given in U.S. Pat. Nos. 5,070,032, 5,095,344, 5,315,541, 5,343,063, 5,661,053, 5,313,421 and 6,222,762. In particular, examples of memory devices with NAND cell structures are described in U.S. Pat. Nos. 5,570,315, 5,903,495, 6,046,935. Also, examples of memory devices utilizing dielectric storage element have been described by Eitan et al., “NROM: A Novel Localized Trapping, 2-Bit Nonvolatile Memory Cell,” IEEE Electron Device Letters, vol. 21, no. 11, November 2000, pp. 543-545, and in U.S. Pat. Nos. 5,768,192 and 6,011,725.

In practice, the memory state of a cell is usually read by sensing the conduction current across the source and drain electrodes of the cell when a reference voltage is applied to the control gate. Thus, for each given charge on the floating gate of a cell, a corresponding conduction current with respect to a fixed reference control gate voltage may be detected. Similarly, the range of charge programmable onto the floating gate defines a corresponding threshold voltage window or a corresponding conduction current window.

Alternatively, instead of detecting the conduction current among a partitioned current window, it is possible to set the threshold voltage for a given memory state under test at the control gate and detect if the conduction current is lower or higher than a threshold current. In one implementation the detection of the conduction current relative to a threshold current is accomplished by examining the rate the conduction current is discharging through the capacitance of the bit line.

FIG. 3 illustrates the relation between the source-drain current I_(D) and the control gate voltage V_(CG) for four different charges Q1-Q4 that the floating gate may be selectively storing at any one time. The four solid I_(D) versus V_(CG) curves represent four possible charge levels that can be programmed on a floating gate of a memory cell, respectively corresponding to four possible memory states. As an example, the threshold voltage window of a population of cells may range from 0.5V to 3.5V. Seven possible memory states “0”, “1”, “2”, “3”, “4”, “5”, “6”, respectively representing one erased and six programmed states may be demarcated by partitioning the threshold window into five regions in interval of 0.5V each. For example, if a reference current, IREF of 2 μA is used as shown, then the cell programmed with Q1 may be considered to be in a memory state “1” since its curve intersects with I_(REF) in the region of the threshold window demarcated by VCG=0.5V and 1.0V. Similarly, Q4 is in a memory state “5”.

As can be seen from the description above, the more states a memory cell is made to store, the more finely divided is its threshold window. For example, a memory device may have memory cells having a threshold window that ranges from −1.5V to 5V. This provides a maximum width of 6.5V. If the memory cell is to store 16 states, each state may occupy from 200 mV to 300 mV in the threshold window. This will require higher precision in programming and reading operations in order to be able to achieve the required resolution.

FIG. 4 illustrates an example of an NOR array of memory cells. In the memory array 200, each row of memory cells are connected by their sources 14 and drains 16 in a daisy-chain manner. This design is sometimes referred to as a virtual ground design. The cells 10 in a row have their control gates 30 connected to a word line, such as word line 42. The cells in a column have their sources and drains respectively connected to selected bit lines, such as bit lines 34 and 36.

FIG. 5A illustrates schematically a string of memory cells organized into an NAND string. An NAND string 50 comprises of a series of memory transistors M1, M2, Mn (e.g., n=4, 8, 16 or higher) daisy-chained by their sources and drains. A pair of select transistors S1, S2 controls the memory transistors chain's connection to the external via the NAND string's source terminal 54 and drain terminal 56 respectively. In a memory array, when the source select transistor S1 is turned on, the source terminal is coupled to a source line (see FIG. 5B). Similarly, when the drain select transistor S2 is turned on, the drain terminal of the NAND string is coupled to a bit line of the memory array. Each memory transistor 10 in the chain acts as a memory cell. It has a charge storage element 20 to store a given amount of charge so as to represent an intended memory state. A control gate 30 of each memory transistor allows control over read and write operations. As will be seen in FIG. 5B, the control gates 30 of corresponding memory transistors of a row of NAND string are all connected to the same word line. Similarly, a control gate 32 of each of the select transistors S1, S2 provides control access to the NAND string via its source terminal 54 and drain terminal 56 respectively. Likewise, the control gates 32 of corresponding select transistors of a row of NAND string are all connected to the same select line.

When an addressed memory transistor 10 within an NAND string is read or is verified during programming, its control gate 30 is supplied with an appropriate voltage. At the same time, the rest of the non-addressed memory transistors in the NAND string 50 are fully turned on by application of sufficient voltage on their control gates. In this way, a conductive path is effective created from the source of the individual memory transistor to the source terminal 54 of the NAND string and likewise for the drain of the individual memory transistor to the drain terminal 56 of the cell. Memory devices with such NAND string structures are described in U.S. Pat. Nos. 5,570,315, 5,903,495, 6,046,935.

FIG. 5B illustrates an example of an NAND array 200 of memory cells, constituted from NAND strings 50 such as that shown in FIG. 5A. Along each column of NAND strings, a bit line such as bit line 36 is coupled to the drain terminal 56 of each NAND string. Along each bank of NAND strings, a source line such as source line 34 is couple to the source terminals 54 of each NAND string. Also the control gates along a row of memory cells in a bank of NAND strings are connected to a word line such as word line 42. The control gates along a row of select transistors in a bank of NAND strings are connected to a select line such as select line 44. An entire row of memory cells in a bank of NAND strings can be addressed by appropriate voltages on the word lines and select lines of the bank of NAND strings. When a memory transistor within a NAND string is being read, the remaining memory transistors in the string are turned on hard via their associated word lines so that the current flowing through the string is essentially dependent upon the level of charge stored in the cell being read.

Sensing Circuits and Techniques

FIG. 6 illustrates the Read/Write Circuits 270A and 270B, shown in FIG. 1, containing a bank of p sense modules across an array of memory cells. The entire bank of p sense modules 480 operating in parallel allows a block (or page) of p cells 10 along a row to be read or programmed in parallel. Essentially, sense module 1 will sense a current I₁ in cell 1, sense module 2 will sense a current I₂ in cell 2, . . . , sense module p will sense a current I_(p) in cell p, etc. The total cell current i_(TOT) for the page flowing out of the source line 34 into an aggregate node CLSRC and from there to ground will be a summation of all the currents in the p cells. In conventional memory architecture, a row of memory cells with a common word line founs two or more pages, where the memory cells in a page are read and programmed in parallel. In the case of a row with two pages, one page is accessed by even bit lines and the other page is accessed by odd bit lines. A page of sensing circuits is coupled to either the even bit lines or to the odd bit lines at any one time. In that case, page multiplexers 250A and 250B are provided to multiplex the read/write circuits 270A and 270B respectively to the individual pages.

In currently produced chips based on 56 nm technology p>64000 and in the 43 nm 32 Gbit×4 chip p>150000. In the preferred embodiment, the block is a run of the entire row of cells. This is the so-called “all bit-line” architecture in which the page is constituted from a row of contiguous memory cells coupled respectively to contiguous bit lines. In another embodiment, the block is a subset of cells in the row. For example, the subset of cells could be one half of the entire row or one quarter of the entire row. The subset of cells could be a run of contiguous cells or one every other cell, or one every predetermined number of cells. Each sense module is coupled to a memory cell via a bit line and includes a sense amplifier for sensing the conduction current of a memory cell. In general, if the Read/Write Circuits are distributed on opposite sides of the memory array the bank of p sense modules will be distributed between the two sets of Read/Write Circuits 270A and 270B.

FIG. 7 illustrates schematically a preferred organization of the sense modules shown in FIG. 6. The read/write circuits 270A and 2708 containing p sense modules are grouped into a bank of read/write stacks 400.

FIG. 8 illustrates in more detail the read/write stacks shown in FIG. 7. Each read/write stack 400 operates on a group of k bit lines in parallel. If a page has p=r*k bit lines, there will be r read/write stacks, 400-1, . . . , 400-r. Essentially, the architecture is such that each stack of k sense modules is serviced by a common processor 500 in order to save space. The common processor 500 computes updated data to be stored in the latches located at the sense modules 480 and at the data latches 430 based on the current values in those latches and on controls from the state machine 112. Detailed description of the common processor has been disclosed in U.S. Patent Application Publication Number: US-2006-0140007-A1 on Jun. 29, 2006, the entire disclosure of which is incorporated herein by reference.

The entire bank of partitioned read/write stacks 400 operating in parallel allows a block (or page) of p cells along a row to be read or programmed in parallel. Thus, there will be p read/write modules for the entire row of cells. As each stack is serving k memory cells, the total number of read/write stacks in the bank is therefore given by r=p/k. For example, if r is the number of stacks in the bank, then p=r*k. One example memory array may have p=150000, k=8, and therefore r=18750.

Each read/write stack, such as 400-1, essentially contains a stack of sense modules 480-1 to 480-k servicing a segment of k memory cells in parallel. The page controller 410 provides control and timing signals to the read/write circuit 370 via lines 411. The page controller is itself dependent on the memory controller 310 via lines 311. Communication among each read/write stack 400 is effected by an interconnecting stack bus 431 and controlled by the page controller 410. Control lines 411 provide control and clock signals from the page controller 410 to the components of the read/write stacks 400-1.

In the preferred arrangement, the stack bus is partitioned into a SABus 422 for communication between the common processor 500 and the stack of sense modules 480, and a DBus 423 for communication between the processor and the stack of data latches 430.

The stack of data latches 430 comprises of data latches 430-1 to 430-k, one for each memory cell associated with the stack The I/O module 440 enables the data latches to exchange data with the external via an I/O bus 231.

The common processor also includes an output 507 for output of a status signal indicating a status of the memory operation, such as an error condition. The status signal is used to drive the gate of an n-transistor 550 that is tied to a FLAG BUS 509 in a Wired-Or configuration. The FLAG BUS is preferably precharged by the controller 310 and will be pulled down when a status signal is asserted by any of the read/write stacks.

Examples of Multi-state Memory Partitioning

A nonvolatile memory in which the memory cells each stores multiple bits of data has already been described in connection with FIG. 3. A particular example is a memory formed from an array of field-effect transistors, each having a charge storage layer between its channel region and its control gate. The charge storage layer or unit can store a range of charges, giving rise to a range of threshold voltages for each field-effect transistor. The range of possible threshold voltages spans a threshold window. When the threshold window is partitioned into multiple sub-ranges or zones of threshold voltages, each resolvable zone is used to represent a different memory states for a memory cell. The multiple memory states can be coded by one or more binary bits. For example, a memory cell partitioned into four zones can support four states which can be coded as 2-bit data. Similarly, a memory cell partitioned into eight zones can support eight memory states which can be coded as 3-bit data, etc.

FIGS. 9(0)-9(2) illustrate an example of programming a population of 4-state memory cells. FIG. 9(0) illustrates the population of memory cells programmable into four distinct distributions of threshold voltages respectively representing memory states “0”, “1”, “2” and “3”. FIG. 9(1) illustrates the initial distribution of “erased” threshold voltages for an erased memory. FIG. 9(2) illustrates an example of the memory after many of the memory cells have been programmed. Essentially, a cell initially has an “erased” threshold voltage and programming will move it to a higher value into one of the three zones demarcated by V₁, V₂ and V₃. In this way, each memory cell can be programmed to one of the three programmed state “1”, “2” and “3” or remain un-programmed in the “erased” state. As the memory gets more programming, the initial distribution of the “erased” state as shown in FIG. 9(1) will become narrower and the erased state is represented by the “0” state.

A 2-bit code having a lower bit and an upper bit can be used to represent each of the four memory states. For example, the “0”, “1”, “2” and “3” states are respectively represented by “11”, “01”, “00” and ‘10”. The 2-bit data may be read from the memory by sensing in “full-sequence” mode where the two bits are sensed together by sensing relative to the read demarcation threshold values V₁, V₂ and V₃ in three sub-passes respectively.

FIGS. 10(0)-10(2) illustrate an example of programming a population of 8-state memory cells. FIG. 10(0) illustrates the population of memory cells programmable into eight distinct distributions of threshold voltages respectively representing memory states “0”-“7”. FIG. 10(1) illustrates the initial distribution of “erased” threshold voltages for an erased memory. FIG. 10(2) illustrates an example of the memory after many of the memory cells have been programmed, Essentially, a cell initially has an “erased” threshold voltage and programming will move it to a higher value into one of the three zones demarcated by V₁-V₇. In this way, each memory cell can be programmed to one of the seven programmed state “1”-“7” or remain un-programmed in the “erased” state. As the memory gets more programming, the initial distribution of the “erased” state as shown in FIG. 10(1) will become narrower and the erased state is represented by the “0” state.

A 3-bit code having a lower bit and an upper bit can be used to represent each of the four memory states. For example, the “0”, “1”, “2”, “3”, “4”, “5”, “6” and “7” states are respectively represented by “111”, “011”, “001”, “101’, “100”, “000”, “010” and ‘110″. The 3-bit data may be read from the memory by sensing in “full-sequence” mode where the three bits are sensed together by sensing relative to the read demarcation threshold values V₁,-V₇ in seven sub-passes respectively.

Page or Word-line Programming and Verify

One method of programming a page is full-sequence programming. All cells of the page are initially in an erased state. Thus, all cells of the page are programmed in parallel from the erased state towards their target states. Those memory cells with “1” state as a target state will be prohibited from further programming once they have been programmed to the “1” state while other memory cells with target states “2” or higher will be subject to further programming. Eventually, the memory cells with “2” as a target state will also be locked out from further programming. Similarly, with progressive programming pulses the cells with target states “3”-“7” are reached and locked out.

Since a verifying take place after a programming pulse and each verifying may be relative to a number of verify levels, various “smart” verifying schemes have been implemented to reduce the total number of verifying operations. For example, since the pulse by pulse programming increasing programs the population of cells towards higher and higher threshold levels, verifying relative to a higher verify level needs not start until a certain pulse. An example of a programming technique with smart verify is disclosed in U.S. Pat. No. 7,243,275, “SMART VERIFY FOR MULTI-STATE MEMORIES” by Gongwer et al., issued 10 Jul. 2007, and assigned to the same assignee as the present application. The entire disclosure of U.S. Pat. No. 7,243,275 is incorporated herein by reference.

FIG. 11 illustrates a conventional technique for programming a 4-state memory cell to a target memory state. Programming circuits generally apply a series of programming pulses to a selected word line. In this way, a page of memory cells whose control gates are coupled to the word line can be programmed together. The programming pulse train used may have increasing period or amplitude in order to counteract the accumulating electrons programmed into the charge storage unit of the memory cell. A programming voltage V_(PGM) is applied to the word line of a page under programming. The programming voltage V_(PGM) is a series of programming voltage pulses in the form of a staircase waveform starting from an initial voltage level, V_(PGM0). Each cell of the page under programming is subject to this series of programming voltage pulses, with an attempt at each pulse to add incremental charges to the charge storage element of the cell. In between programming pulses, the cell is read back to determine its threshold voltage. The read back process may involve one or more sensing operation. Programming stops for the cell when its threshold voltage has been verified to fall within the threshold voltage zone corresponding to the target state. Whenever a memory cell of the page has been programmed to its target state, it is program-inhibited while the other cells continue to be subject to programming until all cells of the page have been program-verified.

Defective Word-Lines

The next sections will consider techniques for the identification of defective word-lines. As discussed in the Background, word-line defects can include both leaky word-lines and broken word-lines. Both of these are consider below, with word-line leakage discussed first.

Word-Line Leakage Detection

Under prior art arrangements, the detection of word-line leakage can typically only be done at test time for the memory chip by applying high voltage levels directly to a device's pins and then measuring the current/voltage levels at the pins. The requires the use of tester device and cannot be done after the memory chip is assembled as part of a device. This means that the word-lines cannot then be checked after device burn-in. The techniques presented here allow for an on-chip means of detecting word-line leakage.

As will be discussed in the following paragraphs, the techniques presented allow for the detection of leakage on a word-line while the word-line has a high voltage applied internally. In an exemplary embodiment, a capacitive voltage divider is used to translate the high voltage drop to low voltage drop that can be compared with a reference voltage to determine the voltage drop due to leakage. The next section will present a related on-chip self calibration method that can help assure the accuracy of this technique for detecting leakage limit. For both of these processes, the can be under the control of the devices state machine, belonging to a built-in self-test to save on the expensive of an external test device. In this way, the leakage determination can be done in an on-chip, automatic process that does not need complicated test equipment and can be performed in the field after chip is packaged.

First, some discussion of the problem involved here is probably useful. There is an ongoing effect to reduce memory devices to ever smaller scales. As the technology scales down to 20 nm and 10 nm memory cells, for example, the distance between the word-lines are consequently 20 nm or 10 nM. Tolerances become more critical and the device is more prone to defects that can cause word-lines leak to the substrate or short to adjacent word lines. It has been found that leakage correlates with dies that fail cycling due to grown defects and that detectable leakage seems to precede actual program status failure.

Previous methods for detection of word-line leakage would force a high voltage on the word-line and measure current leakage from a test pin pad. (Some examples of prior leakage detection is discussed in U.S. Pat. No. 5,428,621.) Since the leakage test requires a very accurate current source, this test mode can only be done by a conventional tester. As manufactures would like to migrate most of the test operations onto an inexpensive tester, a new test flow would be useful to be able to implement on-chip means of detecting word-line leakage. This section presents a way to enable the word-line leakage test automatically and internal to flash memory, and in a way that can be done with various voltage biases and multiple stress topologies. The method can also be done in the field after chip being packaged and further allow to system detect different leakage levels.

For a typical device, the word-line leakage could be on the order 100 nA at high voltage stress such as 10 to 20 Volts. The difficulty of detecting such a small current at high voltage is due to the current NAND architecture. This can be illustrated with FIG. 12. The planes of a memory circuit can have on the order of several thousand blocks, one of which is shown at 610 and each block may have several dozen word-lines, three of which are explicitly shown as WLn−1 615, WLn 613, and WLn+1 611. The high voltage is normally applied on the selected word-line, such as WLn 613 during program and read operations. The NAND architecture also requires to have the least area penalty of the wordline voltage drivers. The driver is typically connected to the wordlines from one end of the wordline array. If the architecture allow the connection to wordlines from both ends, wordline leakage or breakage can be detected by sending a known current from one end and detect the same current from the other end.

The high voltage VPGM is generated by a pump (discussed below with respect to FIG. 13) and supplied to the first decoding CGN block 601, represented here as a switch. CGN block 601 is a block to supplied the various (typically 3 to 5 different kinds) of voltages according to the mode of operations for each global control gate (CG) lines. Three of the CG lines (621, 623, 625) are shown explicitly, corresponding the shown word-lines. The CG lines (as many as the number of word-lines in each block) will rout to the row (block) decoder of the memory array. As indicated by the ellipses, the CG lines run to the other blocks of the array in addition to the only shown block of 610, so that these CG lines usually route with the top metal layer and run through all the row decoders of all planes. In one preferred embodiment, each block is decoded with a local pump. When the block is selected, a logic signal will enable the local pump to apply a high passing voltage transferG on the gates of the passing transistors (here represented by 631, 633, and 635 for the three shown word-lines) in the row decoder. The high voltage on the correspond global CG will be transferred to the word-line of the selected block. Here, only the word-line WLn 613 is shown connected to receive VPGM, with the two adjoining word-lines (611, 615) taken to ground (or more generally the low voltage level), corresponding to the word-line to word-line leakage test pattern discussed below.

During the word-line leakage test, the word-lines can have different bias topology according to the defects to be detected. In the case of detecting word-line to substrate short, all the word-lines can be biased to high voltage of same levels, with the substrate at ground. In the case of detecting word-line to neighbor word-line shorts, the word-lines in the block will be biased alternatively at high voltage (VPGM) and 0 volts, as shown in FIG. 12. The worst parasitic capacitance will be from the latter case.

FIG. 12 also shows some exemplary, estimated values for the parasitic capacitances involved. From the high voltage pump to the CGN (high voltage to multiplexing block) in a 64 word-line architecture the contribution is roughly 5 pF. Inside the CGN block, the loading will be 4 pF. Each global top metal routing from CGN block to the row decoder at the edge of the memory array is 4 pF. The junction capacitance of one plane is 1 pF. Each local word-line has 2 pF.

In the alternative bias configuration, with a total of 64 wordlines, of which 32 wordlines are biased to a high voltage while the other 32 wordlines are biased to 0V, such as shown in FIG. 12, the total word-line capacitance is 2×32=64 pF. The total global CG line will be 5×32-160 pF. To detect the leakage on the high voltage supply node VPGM, then the total capacitance will be 64+160+4+5=233 pF.

Were the system to use 100 nA of leakage to discharge the large capacitance of 233 pF and let the high voltage to drop 1 volt, this will need a wait of 2.3 ms. After detecting the leakage on even word-line, the odd word-line will be tested with another 2.3 ms. The total leakage test time is around 5 ms.

To reduce the detection time, the voltage drop required for the detection can be reduced to 100 mV, with the corresponding detection time reduced to around 500 us. This can be used for in-field detection operations. In one preferred set of embodiments, this could be executed before each erase operation. For example, the detection can either be included as part of the erase operation sequence or can be done before the erase in response to an instruction issued by the controller. \If a block fails, the controller can then remove it from the pool of usable blocks.

The discharge and testing time will depend on the parasitic capacitance of the CG routing. Because of this, one set of preferred embodiments has an on-chip calibration mechanism built in to memory chip so that the precise leakage criteria can be used for detection and the test time can be automatically adjusted according to the chip architecture, word-line voltage stress topology, number of planes, and any other contributing factors. This calibration system is discussed further in the next section.

A normal high voltage pump is normally regulated by a resistor divider, such as shown in FIG. 13. The high voltage VPGM will be divided by the resistors 645 and 647, connected to ground (or more generally the low voltage level) through the switch SW1 649, and the compare point voltage for the amp 643 will be voltage reference vref of usually around 1.2 volts. The resistor chain normally will have a leakage current of 10 uA level. The differential amplifier or comparator 643 will be used to output a digital voltage flag_pump which will be used to control the pump clock. When the pump is pumped to the target level, the flag_pump will be low to turn off the pump clock. When the high voltage is dropped below certain level, the flag_pump signal will go high to enable the pump clock and turn on the pump to supply high voltage.

More detail on charge pumps can be found, for example, in “Charge Pump Circuit Design” by Pan and Samaddar, McGraw-Hill, 2006, or “Charge Pumps: An Overview”, Pylarinos and Rogers, Department of Electrical and Computer Engineering University of Toronto, available on the webpage “www.eecg.toronto.edu/˜kphang/ece1371/chargepumps.pdf”. Further information on various other charge pump aspects and designs can be found in U.S. Pat. Nos. 5,436,587; 6,370,075; 6,556,465; 6,760,262; 6,922,096; 7,030,683; 7,554,311; 7,368,979; 7,795,952; 7,135,910; 7,973,592; and 7,969,235; US Patent Publication numbers 2009-0153230-A1; 2009-0153232-A1; 2009-0315616-A1; 2009-0322413-A1; 2009-0058506-A1; US-2011-0148509-A1; 2007-0126494-A1; 2007-0139099-A1; 2008-0307342 A1; and 2009-0058507 A1; and application Ser. Nos. 12/973,641 and 12/973,493, both filed Dec. 20, 2010, and Ser. No. 13/228,605, filed Sep. 9, 2011. In particular, U.S. Pat. No. 7,554,311 describes a regulation scheme that also employs capacitances in a voltage divider for regulation.

A detection principle similar to FIG. 12 can be used to detect the voltage change on the large parasitic high voltage node. Since the leakage is in the order of 100 nA, a new way to divide the high voltage to low voltage has to be used. A comparator is normally built with a low voltage supply for saving Icc current. A capacitive divider has the advantage of no leakage current.

The difficulty with a capacitive voltage divider is that the initial voltage at the detecting point has to be accurately set. As shown in FIG. 14, a new set of differential amplifiers or comparators 653 is added for the word-line leakage detection on top of that of regulator 643. The comparison voltage verf1 can be set by a digital to analog converter voltage circuit 651, whose input can be set according to the device. (In an alternate embodiment, this could also be set as part of the calibration process.) A switch transistor SW2 659 will be used to initialize the compare nodes at the same voltage level of the regulating level. The capacitors C1 655 and C2 657 are the capacitive voltage divider. A ratio is 1:1 can be used. The detection point voltage Vmid will have a delta of

${\Delta \; {Vmid}} = {\Delta \; {V_{output}\left( \frac{C_{1}}{C_{1} + C_{2}} \right)}}$

where ΔV_(output) is the high voltage drop due to leakage.

To be able to detect the high voltage change of 100 mV, if the C1=C2, then a 50 mV change will be shown at the comparator point. The reference voltage for the comparator will be moved down by 50 mV. If the comparator also has accuracy problems, then the minimum detectable voltage drop will be limited by the comparator. The on-chip calibration can also correct some of the offset and error of the comparator.

The word-line leakage detection is a 3 step detection process, as shown in FIG. 15 where the level on the word-line is shown at 705. In a pre-charge phase, the word-lines are pre-charged to the high voltage level where the pump is on with regulator setting to targeted level SW1=vdd. Enough time should be used to charge the whole word-line. The far side of word-line which located far from the word-line driver side may take a longer time to charge (as shown as the dotted line 707). The high voltage can also pumped in two stages: first pumped to an intermediate voltage with another stronger pump, then use the high voltage pump to charge the word-line to a higher level. During the pre-charge time, the detection point Vmid is also initialized by turning on SW2.

After the word-line is fully charged to the target level, the pump will be turned off (float), along with the resistor regulator (SW1=0). The SW2 is also turned off, trapping a voltage on the mid node.

After some discharge time (a timer can be set with a parameter), the voltage drop will be measured by the comparator 653. The discharge time will depend on the total parasitic capacitance and the targeted detecting leakage current. (For more accurate leakage detection, self-calibration circuits will be introduced in the next section.) The midpoint voltage will be compared with the vref1 to generate the signal Pass or Fail (P/F). The vref1 voltage is generated from an analog voltage generator 651 which can deliver a voltage between 0 to 1.2 V with 50 mV resolution, as an example.

When word-line leakage is detected, the whole block will typically be marked as s bad block which will not be used. Any valid data could be transferred as needed to another block, although, as noted above, in a preferred set of embodiments the leakage detection process is executed as part of an erase process. In other cases, for example when the memory has a NOR architecture, single defective word-line could be mapped out.

On Chip Self Calibration For Detection Time

The word-line leakage detection time depends on the parasitic capacitance, which can have large variations depending on architecture, voltage bias topology, and the number of planes. It is, consequently, preferable to have a method to calibrate the discharge time with a known leakage current. An on-chip self-calibration algorithm is described in this section. A convenient way of accomplishing this, without needing to add extra elements, is to utilize a known current in the regulator to calibrate the detection time.

FIG. 16 shows the same elements as in FIG. 14, but as shown in FIG. 16, the resistor voltage divider is used to discharge the high voltage during the calibration process, as shown by the current path Idis 673. These elements are again preferably implemented as peripheral circuitry on the memory chip and the path tested in the calibration process should match the path actually used for detection of leakage. During on-chip self-calibration, a good block should be used to determine the characteristics of a block without any word-line leakage. The good block may be determined by its program characteristics or from other some other good block check. For example, data corresponding to the highest state can be programmed and read back to see if it is correct. When the calibration is done on a fresh die, word-line leakage will often not have begun to manifest itself and the location of a good block is generally easy. The calibration is similar with the real leakage test and can be performed in 3 stages, as shown in FIG. 17.

A first phase pre-charges the word-lines of the test block to the targeted voltage level pattern by turning on the high voltage pump, the CGN voltage selection circuits and the row decoder for selected block. The high voltage is regulated by the resistor voltage divider and the comparator to enable pump clock. In this step, SW1 and SW2 are both on, as shown respectively at 801 and 803. The word-lines charge up as shown at 805 and 807, respectively corresponding to 705 and 707 of FIG. 15.

The discharge phase will be different from the normal word-line leakage test illustrated in FIG. 15. During the discharge phase, the resistor voltage divider will be kept on with SW1=Vdd. But the pump is disabled and left floating and SW2=0 to isolate the mid node from the resistor divider. The high voltage VPGM will be discharged through the resistor chain with a fixed leakage current along the path 673 of a discharge current of I_(dis) on the order of 10 μA.

When the output P/F 809 of Diff Amp 653 flips after comparing with a selected vref1 value, the amp output Pass/Fail will feed back to turn off SW1. A timer can start counting the time from the start of the discharge phase till the comparator flipping of P/F from pass to fail.

Based on detecting leakage-detection criteria and the ratio of this to the resistor leakage, the timer can be multiplied by a factor of 2 (such as 128) to set the timer counter for detecting targeted leakage current. For example, if the resistor leak 10 μA, the timer multiplying 128 will give the detecting current of 78 nA. (Other factors could also be used, but factors of two are readily implemented, as an easy way to multiply by 2 is to perform a shift of binary digits to the higher bits.)

The calibration only needs to be done once for a given voltage topology during die sort test. The timer digits can then be fixed and stored, for example, in a ROM fuse block. During power on read, the timer digits will be read out to registers and controls the word-line leakage test. For a different stress topology, a new calibration is needed, since the parasitic capacitance is changed. After each calibration, a corresponding timer parameter can be acquired and saved in the ROM flash memory.

The word-line leakage can be used during manufacture test, or for in-field tests once the device is out of factory. The micro-controller would issue the command to do the word-line leakage test in the user application. A convenient time to do the leakage test is before the erase operation, since the program disturb incurred during the leakage test can be eliminated by the subsequent erase operation.

Detection of Broken Word-Lines

This section looks at the detection of broken word-lines. As device size decreases, in addition to the likely increase in leaky word-lines, the occurrence of broken word-lines is also likely to become more common. A broken word-line will have a high resistive connection across the break, because of which the cells on far end of the word-line (on the other side of the break from the word-line driver) will see a voltage drop during both program and verify operations. This will lead to programming pulses having a lower amplitude, so that cells will be programmed less; but as the verify level is also lowered, these under-programmed cells may still verify. As a result, the threshold voltage distribution for the broken word-line will show two humps, one corresponding to cells one side of the break and the other corresponding to cells on the other side of the break. The method described in this section can be used to identify the broken word-line failure and recover the data of the broken word-line

There are various ways by which the broken word-line failure could be detected. One approach is to use a smart verify scheme, such as is described in US patent publications numbers US-2010-0091573-A1 and US-2010-0091568-A1. In this arrangement, the program voltage level is recorded when a certain number of bits pass the lower page program operation on each word-line. This recorded program voltage level is then used as a starting program voltage for the upper page of the same word-line. With this scheme, the number of program loops for each word-line is largely uniform, hence any variation in the total program loop number may be used as an indication of a broken word-line. However, as the program loop number in a broken word-line may not be significantly higher than typical, using the total program loop count to judge this failure could result in false alarms

Another approach to detect this sort of failure is the “forbidden zone” read, where a read is performed to determine whether any cells have a threshold voltages in the region between the ranges allotted to data states. (See, for example U.S. Pat. Nos. 7,012,835; 7,616,484; or 7,716,538.) In this kind of scheme, after the program operation completes, a particular state can be sensed at two different levels and the results of the two sensing operations can be compared with each other. A scan operation can then be done to check then number of bits between the gaps of two reads which were sensed as non-conducting in one sensing operation, but conducting in the other sensing operation. This solution comes with performance penalty as every single program operation would be followed by two read operations and a scan operation

Yet another method of identifying broken word-lines is to screen out the failure during die-sort. In this method, a whole block is programmed and then read back. (For example, when data is stored in a multi-page format, the lower page of each word-line can be programmed and read twice.) One read is done with a normal read point and another with a raised read point, similar to a forbidden zone read described in the last paragraph. The results of the two sensing operations are then compared using a test-mode command sequence. However, this will only pick up the word-line breakage that manifests itself at test time, when the symptoms often do not show up until the device has operated over some time. Also, when the word-line already exhibits breakage, it may not demonstrate this on every program cycle and, consequently, may be missed in a single test operation

Considering the problem further, the symptom of broken word-line failure is a distribution with two humps. FIG. 18 shows the threshold distribution of a block of a memory word-line by word-line, for the a 64 word-line example. The distributions for three states are shown at 901, 903, and 905. As shown, these form three well defined and separated humps, where the highest two states, for example, are separated by the region between 923 and 925. For a broken word-line, however, those cells on the far side of the break from the word-line driver will be shifter to lower threshold values, as shown at 911, 913, and 915.

The reason behind a double hump distribution is that the part of word-line at far end of the word-line driver will show voltage drop. As a result, the cells that are located at the far end of the word-line will program slower and pass verify at a lower voltage. Since the failure will not cause a program status failure, it may not be detectable for a typical program failure mechanism. Programming a broken word-line will show some program loop variation, but word-line-word-line and block-block variation make it difficult to judge the failure based on the program loop count, as can be illustrated with respect to FIG. 19. FIG. 19 shows the number of pulse-verify iterations, or loop count, for each word-line to program, in this example, lower page into a 64 word-line block. As shown there, the loop count fluctuates over the different word-lines by several counts. These variations can reflect fluctuations due to the design particulars, such as whether it is an edge word-line or a central word-line, or how many erase-program cycle the word-line has experienced, as well as process variations. In the case of WL50, the loop count is noticeable higher than the other fluctuations, indicating what may likely be a broken word-line, although further tests would be used to confirm whether it is actually broken or this is just a false alarm.

The techniques presented here make it possible to detect broken word-line failure by comparing the program loop count for the cells located on two different sides of the fault. The cells along word-line are programmed and it determined how it takes the cells of different groups or subsets of these cells to verify as programmed to target state, such as writing all the cells to have a programmed lower page. A group with cells on the far side of a break from the word-line driver will take longer to program than a group that has all of its cells between the driven and the break. As memory cells are typically programmed using an alternating pulse-verify algorithm, this can be done by keep track of the number of pulses, or loop count, needed by the different groups or just the difference in the number required. The programming can be done for all of the cells along word-line or some portion of them, such as for system that program the odd bit lines and even bit lines separately. In the exemplary embodiments, the subsets of cells that have their loop counts compared are the contiguous subset of cells of the segment of one end of the word-line and the segment at the other end of the word-line. More generally other subsets of the cells could be used, but by looking at segments from the two ends of the word-line any break should be caught without having to do multiple comparisons of groups' loop counts; and looking at segments of the word-line is generally more readily implementable in the exemplary architecture than if the groups are formed from non-contiguous subsets of the cells, overlapping subsets, or some combination of these. To be able to compare the loop counts meaningfully for the different segments, their cell should be programmed with the random data, for example, in a multi-page format, The loop count comparison between two end of the word-line will eliminate the word-line to word-line or block to block variations. The cells on the same word line will follow similar programming characteristics.

Memory devices often already include a scan to check for failed memory bits when programming. The exemplary embodiment incorporates the broken word-line detection into such a routine, which can have several advantages. One is that such scans may already keep track of the loop count for the memory cells or segments as part of their algorithms. Also, as allows the broken word-line check to be performed many times after the device has been in operation, it can pick up breakages that only manifest themselves after device test or that are not detectable at every test.

In an exemplary algorithm, the broken word-line detection is incorporated into a failed bit detection that is done during the last few program loops and which counts the failed bits segment by segment, the word-lines being subdivided into multiple segments. In the exemplary memory embodiment presented above, each the segments can be taken to correspond to one or several adjacent ones of the read/write stacks as shown in FIG. 7. While this scan is ongoing, the scan result of first physical segment and last physical segment on the ends of the world-line can be monitored. When the failed bit count for either one of these two segments end goes below a fixed (in this example) criterion, a signal is latched high to mark the passing of the one segment.

An up-counter can then be triggered when the first of these segments passes the scan. The counter is then stopped when the slower of the two segments passes scan operation. At the end of program routine, the output of the up-counter is compared to the fixed criterion. If the count is higher than the criterion, a signal can be latched high to indicate that a broken word-line has been detected. The up-counter can be implemented on the state machine (112 FIG. 8). As the up-counter can simply count the program look stating when the one segment passes the its write criteria, the on-chip state machine will typically be able to keep count of the program loops, so this adds an additional count for it to maintain.

If a broken word-line is detected, its program status should be set to fail and the corresponding cached data should be terminated. The controller can then toggle out the next page of data that if it has been already loaded in the data latches. The next page data can also be programmed to a different location instead of toggling the data out to controller. The data of the failed data page and any corresponding lower pages can then be recovered by issuing a command sequence that will trigger read operation with shifted read voltage levels. (Aspects of data recovery and corresponding latch structures are described in U.S. Pat. No. 7,345,928.)

The process can be illustrated by the diagram of FIG. 20 that shows the waveforms for some of the signals involved in this scheme. In this diagram, OPC is the program loop count, corresponding the iteration in the pulse-verify sequence. OPC_DIFF is the up-counter for counting the program loop difference. SEG1_COMP is the latched signal to indicate the passing point of first of segments. LASTSEG_COM is the latched signal to indicate the passing point of the last segment. FIG. 20 picks up the program process after n−1 loops have been completed at time t₀.

Initially, SEG1_COMP, LASTSEG_COM, and the BROKEN_WL signals are all low and the up-counter is initialized to 0. At t₁, corresponding loop count n, a first one of the end segments (here taken as the first segment) reaches its passing point and SEG1_COMP goes high and the up-counter starts, as shown as OPC_DIFF. OPC_DIFF continues to increment up with the loop count until the other of the end segments (here the last segment) passes at t₄, corresponding to loop count n+3. The signal BROKEN_WL then goes high when OPC_DIFF>F_OPC_DIFF.

One complexity that can involve in implementing the above described scheme is the case when the architecture use two sided word-line drivers, placing drivers on both sides of the array (such as would be in the row decoders 230A and 230B of FIG. 1). This can be illustrated by FIGS. 21A and 21B. In FIG. 21A the word-line WL 901A has the driver 905A to the left, closest to the driver. The last segment along the word-line 901A is on the other side of the break 903A from the driver 905A and will consequently see lowered voltage levels and be slower than the first segment. In FIG. 21B the word-line driver 905B is to the right and closest to the last segment close to the last segment along word-line 901B. In this ease, the first segment will be on the far side of the break 903B, receive lowered voltages and the first segment will be slower than the last segment. Under such an arrangement, it cannot be assumed that the last segment will pass last, since it may be closest to the driver.

The incorporation of broken word-line detection into a failed bit scan routine is considered further for the case where the memory array includes a number of redundant columns (for use replacement of defective columns), which are placed to the left side of the array so that they all are found in the last segment. (Such an arrangement is described in more detail U.S. Pat. No. 7,170,802, for example.) One way of implementing a failed bit scan for such a circuit is to scan the segments in the following order: Nth segment (last segment)−1^(st) segment−2nd segment . . . (N−1)st segment. The Nth segment is checked first since this will give an indication of the number of available spare columns to which data from defective columns in other segments can be remapped. In a normal segmented bitscan, such as that described in U.S. Pat. No. 7,440,319 and which can serve as a basic embodiment upon which this discussion can build, if one segment failed the criteria, the rest of the segment will not be scanned to save time. If segment N fails, the circuit does not proceed to scan the first segment. The process then moves through the other segments, where the criteria for these other segments will preferable consider not only the number of failed bits in this segment but also the number of failed bits in the last segment counting the failures of the replacement columns. In an exemplary embodiment, in case of two sided word-line drivers, the scan circuit should be modified such that it continues to scan the first segment even if the last segment fails. This is shown in FIG. 22. Under this exemplary embodiment, the segmented bitscan is include as part of a normal program algorithm. The broken word-line detection scheme can be integrated into a similar state machine to that for segmented bitscan without the detection. In this arrangement, when the last segment fails, the segmented bitscan is not terminated because the first segment should also be checked to see at which program loop it passed the program. Under the arrangement of U.S. Pat. No. 7,440,319, if any segment does not finish programming, it is counted as the whole page of data not being finished, and terminates as soon as one segment fails. Similarly, when the broken word line detection is incorporated, the exemplary embodiment will go through the last segment and continue to the first segment regardless of whether the last segment fails or passes.

In addition to the changes just described with respect to FIG. 22, the failed bit scan routine is also modified to include the broken word-line detection process. As noted above, the exemplary embodiment includes redundant columns, so the number of failed bits allowable in the other sections depends not just on the number of failures there, but also on the number of redundant bits available in the last segment that could be substituted in for failed bits in the first segment. For example, the failed bit count of last segment and first segment are added together and then compared to the criterion in order to determine pass/fail status for first segment. In the exemplary embodiment that incorporates broken word-line detection, the flow will be modified such that the failed bit count for failed segment can be compared to the failed bit criterion in order to determine pass/fail for first segment. FIGS. 23A and 23B show the comparison between a counting scheme that includes neither broken word-line detection nor the process of FIG. 22 (FIG. 23A) and the exemplary embodiment that includes both (FIG. 23B).

The first of these is schematically illustrated in FIG. 23A, which starts at 1001 with a scan of the last segment, including the redundant columns (ColRD), which is then compared against its criteria at 1003 to determine if the last segment has failed. In this embodiment, the process continues (pass or fail) on to the first segment scan at 1005. The criteria used at 1007 for the first segment is compared not just to the scan result for the first segment itself, but also takes into account the number of redundant columns (ColRD) available. If the 1st segment test at 1007 is passed, the flow similarly continues on to the second segment at 1009 and 1011, and so on through the other segments.

In order for the scheme to work correctly in case of two sided word-line drivers, the scan circuit will need to be modified such that it continues to scan the first segment even if the last segment fails. The diagram of FIG. 23B shows a scan algorithm to account for this and that includes the broken word-line check. As before, the scan of the last segment 1051 is compared against the corresponding criteria at 1053. In this embodiment, the process will again continue on the scan of the first segment, 1055, regardless of whether or not the last segments passes or fails, going from 1053 to 1055 if 1053 fails. If 1053 passes, the flow will now go to 1059 as well as 1055. It should be noted that a broken word-line does not necessarily fail to program When the segment is far from the word-line driver, it will be slower to program, by not necessarily impossible. Hence, it may eventually pass, but it is needed to determine the programming speed at both ends of the word-line, which may differ significantly, in order confirm that a word-line is actually broken.

When the last segment passes, it will trigger the OPC_DIFF block, as will the first segment from 1057, with the first of these to pass starting the counting and the last to pass stopping it in order to count the difference. At 1057 it is judged whether the first segment itself, without the inclusion of redundant columns passes or fails. As noted, the determination of word-line breakage at 1059 will be based difference from the first segment (alone, without redundant column considerations) and last segment loop counts. 1061 is the bitscan for program status as before, where columns of the first segment may have defective columns replaced by redundant columns (from the last segment). Because of this, both 1057 and 1061 are included in the flow. The process then continues on to the second segment at 1063, 1065 and other segments as before.

By introducing this scheme, the number of defective devices due to broken word-line failures can be reduced without performance penalty. Further, as this is included as part of the programming routine, it is able to pick up breaks that only manifest themselves after a device is shipped. This allows it to be a more efficient and accurate method of broken word-line detection compared to the other methods due to the fact that it is in-field detection. It can reduce the program loop count variation due to word-line-word-line, block-block and chip-chip variations with no performance penalty and avoids time-consuming die-sort screens.

Determination of Word-Line Leakage by Current Comparison

This section returns to the consideration of word-line leakage and considers some additional techniques for its detection. As before, the techniques can be used to detect word-line leakage to the substrate or to neighboring word-lines. The techniques discussed in the following can be particularly useful for distinguishing the presence of leakage current against the background noise current of a device. Even when no word-lines are selected, there will be some junction leakage current on the path by which the word-lines receive their voltage as it is delivered to through junctions connected to supply the control gate voltages. For example, referring back to FIG. 12, even when the word-line select gates 631, 633, 635 are turned off, there will be some leakage. This background leakage noise will vary with process and temperature variations, and can range over a rather large difference of magnitude. To remove this background current noise, the techniques of this section compare the current drawn by a plane of the memory when no word-lines are selected to the current drawn when a set of word-line are selected. This can be done by using two different planes or by the same plane, where one of the values (such as the reference value with no word-lines selected) is measured and recorded and then compared to the other value. As to when the testing process is performed and the patterns of word-lines selected for leakage testing, these can be the same as discussed above with the other embodiments on detection of word-lines leakage.

The concept can be illustrated with respect to FIG. 24 for the case where two different arrays are used, one unselected and one with an erase block selected for testing, and the two currents are generated at the same time and compared directly. A voltage level, in this example 20V, is generated in a supply VSUP 2021 and supplied through respective resistances R 2023, 2025 to the word-line decoding circuitry of a pair of planes (semi-autonomous memory arrays) Plane 0 2001 and Plane 1 2011, each taken to be of the same design of 2K erase blocks and having respective word-line decoding circuitry, including the word-line selection switches, XDEC of 2003 and 2013. In Plane 0 2001, Block 0 is selected for leakage testing. To do this, the selection circuitry applies a voltage pattern to Block 0's word-lines; for example, a stripe pattern could be applied with the high voltage from line 2005 on ever other word-line and the non-selected word-lines left to float or taken to ground by line 2007. This will detect leakage between the word-lines and also between the (high) word-lines and the substrate. The word-lines in the unselected blocks of Plane 0 2001, and all of the blocks in Plane 1 2003, are then left to float with their word-line select switches turned off. Even though Plane 1 2011 is non-selected, all of the same voltages are supplied along lines 2015 and 2017 to Plane 1's XDEC circuit 2013.

Even though no word-lines are selected in Plane 1 2011, it will draw a certain amount of current (I₁=I_(junction)) due to junction leakage that will serve as a reference value. Plane 0 2001 will draw both junction leakage current and any word-line current I_(leak). By looking at the voltage difference between node N₀ and node N₁, the current leakage can be isolated. If the voltage difference, ΔV=(V₁−V₂)=I₁R−I₂R=I_(leak) R, as compared in COMP 2027 exceeds a threshold, the Detect signal is asserted to indicate that the selected Block 0 has leaky word-lines. In response, the bad block can be mapped out or other corrective actions taken.

Consequently, by using a reference plane and current sensing, the word-line leakage can be detected. Any background noise (junction leakage) will be cancelled through the comparison, since both sides see the same amount of junction area. There is then no reference noise due to differential sensing. Also, the sensing speed is improved since it uses current sensing; and, much as described further for the embodiment presented below, additional reference current can be injected in reference path for margin test.

The arrangement of FIG. 24 requires a device with at least two planes. To be accurate, it also needs that the planes are well-matched: that the reference and selected planes and word-lines have good matching in loading; that the background junction leakage average of the 2K blocks should be at the mean of the distribution; the leakage current on the selected and reference word-lines should match; that the actual leakage criteria is measurable over the difference background leakage; and that the device and resistors are well matched. Also, as the current being drawn through the resistance 2023 is increase by higher I_(leak) values, this can affect the value of R and result in the I_(junction) component not accurately cancelling between the planes (although this effect can be reduced by use of current mirrors, much as described below). The next embodiment overcomes or reduces these various complications.

More specifically, in one preferred embodiment word-line to word-line and word-line to substrate leakage can be detected in-field by determination of a reference level, with all word-lines and blocks de-selected, and then applying a stress mode level done on the same plane by applying voltage levels in a stripe mode on a selected set of word-lines, typically taken to be from a single erase block. To facilitate the accuracy of this process, the exemplary embodiment uses a current mirror scheme.

To give an idea of the current levels involved, calculations for a typical device (based on electronic design rules) give values for maximum junction current at high temperatures to be on the order of several micro-amps and at low temperatures to be on the order of a few tens of nano-amps. Thus, the level of background noise from the junction leakage can vary widely depending on conditions. The amount current that a leaky word-line, whether the leakage is to another word-line or to the substrate, be on the order of a hundred nano-amps to tens of micro-amps. As these numbers illustrate, the relative amount of noise (junction leakage) can be high when compared to the signal (word-line leakage), and both values can vary significantly depending of operating conditions. Consequently, to be able detect word-line leakage, it is useful to have a common-mode current to speed up the detection time, particularly in case junction leakage is low.

In the exemplary embodiment, each plane independently uses a current mirror and current sensing to determine the word-line leakage current. To determine leakage, current is sensed two times (a reference current value and word-line leakage current) and the values are stored digitally and then compared at the end of operation. This arrangement helps to reduce the error terms from current mirror, leakage, and op-amp mismatch. The basic operation can illustrated with respect to FIG. 25.

The current being drawn by the array, where for determining the reference value or for the leakage test, is mirrored and flow down the left side through the transistor 2105. The level MON is at a high voltage and FLG, on the other side of the inverter 2101 is low. The gate of the transistor is controlled by the 8-bit (in this example) DAC counter 2107 which starts from the high value (FF in hex) and decrements downward. (Alternately, it could start at the low end and increase, but starting high can have benefits in terms of settling times.) The count continues until the level at MON is pulled above a trip point of FLG, at which point the counter value is stored into the latch Register 2103. After doing this once to determine the reference values, the selected testing pattern is then applied and the process repeated, after which the results are compared to determine if the leakage for the selected set of word-lines exceeds the allowed amount. (Alternately, the reference level determination could be second.) If multiple blocks, or differing applied voltage patterns (e.g., switching which word-lines are high and which are low) are to checked in the same set of test, the reference level need only be done once and used for the various comparisons.

FIG. 26 shows the elements from FIG. 25 along with some to the other elements used in the exemplary embodiment for the leakage determination circuitry. These elements are connected between the charge pump circuitry 2240, which supplies the high voltages used to apply the test voltages, and the decoder circuitry of the memory plane or planes, which applies the voltage to a selected set of word-lines. In the exemplary embodiment, this circuitry is split into a part that is specific to each plane and, for multi-plane devices, a portion that is shared between multiple planes. The example uses a 2-plane device.

The charge pump system UMPUMP 2240 includes the pump 2241 itself that supplies the high voltage used for testing (here a programming voltage VPGM) and also the resistance 2249 and the comparator 2251 used to set and regulate the value of VPGM. The transistors in between will be discussed below. The voltage is then supplied to the leakage determination circuitry 2220 and 2230.

The portion specific to plane 0 is 2220-0 and the portion specific to plane 1 is 2220-1, with the shared portion as 2230. The portion 2220 transfers the voltage form the charge pump system UMPUMP 2240 to the decoding and word-line circuitry VCGSEL 2260-0, here shown only for plane 0. This is supplied as VCGSEL_P0 and the switch of high voltage transistor 2211 is used to by-pass the detection circuitry for normal operations by passing VPGM to the gate of 2211 by closing the circuit through LVSH. Similarly, during normal operations, the high level of LVSH is also applied to the gate of the high voltage switch of transistor 2243 and the pump system UMPUMP 2240 operates in its typical manner by-passing 2245 and 2247. During testing, both of 2211 and 2243 will be off.

During testing, to determine the amount of current being by the plane, the amount of current being drawn needs to be determined using the process described above with respect to FIG. 25. Rather than use the actual current, and thereby dilute but the measuring process, the current is mirrored by the current mirror 2213, here formed by two pairs of high voltage PMOSs. As some voltage is dropped across the path of the mirror 2213 between UMPUMP 2240 and VCGSEL 2260-0, the two high voltage PMOSs 2245 and 2247 are used to replicate threshold voltage drop across the mirror.

The mirrored current is then used to detect the leakage, whether the reference value or the actual leakage test value, as described with respect to FIG. 25. Transistor 2205, comparator 2201 (here just represented as an inverter), counter 2207 and register 2203 of FIG. 26 respectively correspond to the elements 2105, 2101, 2107 and 2103 of FIG. 25. The comparator 2201 is now also explicitly shown to have as input a reference voltage Vref that could be supplied by, for example, a bandgap circuit. The capacitor 2209 is also added to smooth out the VMON level. In this embodiment, the elements 2205, 2209, and 2201 are specific to block 2220 of each plane. The block 2230, including counter 2207 and the register/comparator 2203, is shared between block. The counter/comparator 2203 will have a register for each block (2 by 8-bits in this example) and one comparator that can be shared by both planes.

Block 2220 also includes Icm 2215 as a common mode current source to set a minimum current flow through the current mirror to meet settling/detection time. The offset detection current source Ioff 2217 is used during the leakage determination process (including determining the reference value) so that a good block is not detected as bad due to noise determination or detection of error. The offset detection current is used to set a threshold to mark bad block for this purpose.

FIG. 27 is a block diagram to schematically illustrate the relationship of the elements of FIG. 26 to the two planes. The planes P0 2270-0 and P1 2270-1 each have their respective word-line decoding circuitry VCGSEL 2260-0 and 2260-1 that includes the word-line select switches (631, 633, 635 of FIG. 12), as well as any other intervening selection circuitry. In addition to selectively applying VPGM to word-lines, this will apply any other appropriate word-line levels for the various memory operations, where these voltages are collectively indicated as Vcg. (The other peripheral circuitry of the memory circuit is again suppressed here to simply the discussion.) The blocks 2220-0, 2220-1, 2230, and the pump UMPUMP 240 are then as discussed with respect to FIG. 26. In normal programming operations, the blocks 2220-0, 2220-1 are basically by-passed and the same basic paths can be maintained, delivering VPGM as needed; during the word-line stress, however, the new features are then used.

FIG. 28 is a timing diagram for one particular implementation. Here the reference value for the array, but the order could be switched. In this first sub-operation, only the load from the word-line decoding and selection circuitry (or XDEC) is driven to determine the reference leakage. First is an initial ramp-up and stabilization time, during which the current mirror circuits (2213, and 2245, 2247) are bypassed. Here the VPGM value is taken as to be around 17V and the duration of this phase could be something like 50 micro-seconds, to give them concrete values. Next, the level of VPGM is stepped up to compensate for the voltage drop across the current mirror by replicating the threshold voltages along the path. The LVSH switches are disabled on the current mirror circuits are enable. This takes VPGM to something like 17V+2 Vtp, where Vtp is the threshold voltage of the PMOS devices in the path, so that VPGM is ˜20.5-21V. This leave VCGSEL, on the other side of the transistors, back at around the initial VPGM value of ˜17-17.5V. Here, 200 μs are allotted for this sub-phase. After this follows the detection and store data sub-phase. The duration of this part is determined by the step time of the counter's (2207) time step. Once the reference value is set, then VPGM can be taken low and the leakage determination phase can begin.

To detect the word-line leakage, the load will now include a selected set of word line as well as the decoding circuitry. The first two sub-phases are largely the same, except of the change in load. The last sub-phase, of detection and latching the value is much the same, except that it will now also include the comparison of the two results and the determination of whether the selected set of word-lines, typically a block, contains any leakage. If any additional checks are to be done at this time, such as for instance switch the selected pattern between the even and odd word-lines of a block, they can be executed using the same reference value.

FIG. 29 repeats the waveform for one (it can be either one) of the phases of FIG. 28, but also shows the leakage current from the array superimposed with the stepping current as determined by the counter (2207, FIG. 26 or 2107, FIG. 25), and the FLG signal from the comparator 2201. Once VPGM is take high in the ramp-up and initialization phase, any leakage current begins to be drawn and the VPGM value is then stepped up. For the detection, the stepping current then begins counting. In the exemplary embodiment, the count will start high and then decrement as this will allow better VPGM settling time and better accuracy through the detection range. Once the stepping current falls below the leakage current, FLG goes low and the data value is latched in register. Once both the reference (REF) and leakage detection test (DET) values are latched, then can be compared. If DET>REF, the block (or other selected set) is marked as bad, and if DET<REF, the block is marked as good.

FIG. 29 illustrates the “normal” case, where the leakage is within the range of the stepping current. The logic also preferable handles the cases where the leakage exceeds the stepping current range and also when the level of leakage is very small. For the case when the reference leakage is high, the result is marked as good, while for leakage determination test, the block will be marked as bad. The case of the leakage current being less than stepping current for the REF value determination can be avoided as the common mode current, offset current, or both can adjusted to set the REF leakage to lie in the stepping range. For the DET value, if this is below the stepping range, the block (or other selected set of word-lines) is marked as good.

Current Determination Based on Charge Pump Regulation Clock Charge Pump Based Over-Sampling ADC for Current Detection

This section considers a further technique for determining leakage in a word-line based upon the charge pump system's behavior while driving the word-line under regulation. This technique is more broadly applicable for determining load current or leakage current from other circuit elements under charge pump output bias and, even more generally, for determining the magnitude of current values supplied from the output of charge pump. Relative to the arrangements of the preceding sections, in many applications this charge pump-based approach can have a number of advantages, including speed of operation, reduced susceptibility to noise, and lower power consumption. It can be executed while executing other on-going operation, rather than as a separate process. Further, as it requires little additional circuitry beyond the standard regulatory elements of charge pump system, it is readily executed for devices that already include a charge pump.

FIG. 30 illustrates an underlying basis of the process used in this section. When a charge pump is driving a load in regulation, then I_(output)=I_(load), determining simple relation between pump load (or leakage current) and the pump clock signals used by the pump to maintain regulation: I_(output) (pump)=C*V*f, where C is the pump coupling capacitance per stage; V is the clock amplitude driving the capacitance C; and f is the frequency of pump clock to maintain the output load of the pump I_(load). Both C and V are typically fixed values for a pump design, with the frequency varying linearly as load changes. In another word, within a set amount of time T, the number of pump clocks that occur will have linear relationship with I_(load).

FIG. 30 shows this sort of highly linear relationship for an actual device, with a linear relationship of (100 pulses/10 μA) within 50 us. This behavior allows for accurately determining even very low current levels based on counting the number of clocks. This technique of pulse counting for charge pump within a fixed duration incorporates oversampling ADC techniques. Rather than directly comparing the load current with reference current by use of an analog comparator, the pulse counting incorporates many counts over the integration time, and the sampling noise (or background noise) is lowered by accumulating and averaging the results over many pulse counts. If the counting was done only a few pulses, the error could be large due to noise. However by counting many pulses, the randomness of noise is being averaging out over the integration period. The noise floor of current sensing error is reduced through this over-sampling technique.

As the pump is already driving the circuit element connected as load, the detection can be performed in background while the operation goes on, without need for a dedicated leakage detection operation. This arrangement can significantly improve detection accuracy and speed, as well as reducing design complexity.

The discussion below will focus on the case where the charge pump system is driving a word-line on a flash memory array and is used to determine whether that word-line has any leakage, but can used more generally. As it is based on the number of clocks in an interval, it can be used for negative voltage charge pumps as well as the more usual positive voltage case. Thus, as well as word-lines or other elements on a non-volatile memory circuit, it could also be used to detect background leakage on DRAM devices, for example.

The general process can be considered further with reference to FIGS. 31 and 32, which illustrate a pump driving a current load I_(load) and a load of twice I_(load). In FIG. 31 the pump 3001 receives an oscillator signal from the regulation circuitry (not shown) that is based on the output level. The pump 3001 then generates an output current I_(output) that drives the load 3003 at I_(load). While operating under regulation, the number of clock pumps is counted over a time T_(duration) when a control signal Detection Phase is high. In this example, the number of clocks is 6. FIG. 32 schematically illustrates the same situation, but where the load 3003′ now draws twice the current, 2×I_(load), where now 12 clocks are counted during the interval. This count can then be compared to a reference, such as could come from a calibration process or driving other components, to determine that amount of current and whether there is leakage.

FIG. 33 is a block diagram to illustrate further the circuitry involved for typical implementation. The output Vout is fed back to the regulation circuitry 3005, where it is compared to a reference value. Based on the comparison, the signal OSC is then varied to maintain the output in regulation. The OSC signal is also sent to a current detection block 3007 where, when the number of clocks is counted for T_(duration) in response to a control signal Detect. (Here, this is shown as an input to the block 3007, but this could also be determined internally in other embodiments.) The current determination block 3007 then puts out the result I value, which can be presented in different ways according to the embodiment. For instance, if block 3007 is only checking for leakage, the output may just indicate whether or not the count exceeds a reference value. In other cases, the output could be the count, which can then be converted to a current value elsewhere, or the conversion could also be done in 3007. (More detail on charge pump systems and their regulation can be found in the group of references cited above.)

Consequently, this simple relation between leakage, or more generally load current, and the number of pump clocks allows for accurate current determination. Although higher order corrections could be introduced, the high degree of linearity means this is not needed in most applications. As the number of clocks over the interval allows an accurate determination, this can remove the need to insert any additional circuitry in the current path in order measure current. Just need count and make determination. For the exemplary embodiments most of concern here, such as in memory operations, by accumulating the total number of pump clock during a given detection phase, and comparing this with a criterion, which can be predefined or pre-calibrated, the system determine whether leakage is occurring—or, more generally, the amount of load current being drawn. The decision can easily tell if something, such as leaking word-line, is causing an extra load current for the pump.

Returning back to FIGS. 26 and 27, this means that according to the embodiments of this section the elements of blocks 2220 and 2230 are not needed. The pump element 2240 can then be a standard pump system, but with the addition of the current detection elements based upon the count.

FIG. 34 is an exemplary timing diagram for embodiments of this section and corresponds to FIG. 28 of the previous section. Initially, the pump needs to ramp up to, in this case, the VPGM value of ˜47V. Note that unlike in FIG. 28, the level does not need to be pumped to the higher level, but only the level actually used in the operation. Without the word line load attached, the system start checking background noise due to junction leakage and other current drawn even when the component to be measure is not yet connected as a load. Once under regulation, the system acquires the pump clock counts of N. In this example, the different phases are taken to have a duration of 10 us, but other values can be used, including different values for t different determinations. With the word line connected, once out of the recovery phase the system can start to check for the noise leakage plus any word-line leakage current. In the measurement (here 10 us) window, the system acquires pump clock counts of N+M that can then be compared to a reference value to determine whether the word-line is leaking. For example, the system can determine the word line leakage pulse counts M, subtract, say, two counts, and compare it with a pre-defined criterion and make the judgment on if the word-line is leaking. Note that this process can be done more quickly that that shown in FIG. 28 since it does not require stepping up the VPGM values to the higher level before the detection interval and also the big drop in the middle is removed; and as already noted, this can be done on the fly during a regular operation.

Consequently, in many applications the techniques of this section can have a number of benefits relative to those of the preceding section. For one thing, it has lower susceptibility to noise. As the elements 2220 and 2230 of FIGS. 26 and 27 are no longer needed, noise from elements such as the current mirror 2213 of FIG. 26 is eliminated. Also, as a longer integration time can be used to determine the count in FIG. 34 relative to the detection time of FIG. 28, other noise (such as from word-line to word-line coupling) is smoothed out due to an effective over-sampling technique for a DAC encoding of the load's current.

As discussed above, the current from the pump is given by the relation I_(output) (pump)=C*V*f, where C is the pump coupling capacitance per stage; V is the clock amplitude driving this capacitance C; and f is the frequency of pump clock. In the example of I_(output) (pump)=C*V*f, where C is the pump coupling capacitance per stage; V is the clock amplitude driving capacitance; and f is the frequency of pump clock for the output load of the pump I_(load) in regulation. To determine the current being drawn by the load, the number of clocks of a time period is counted; for example, in FIG. 31 6 clocks are counted over the interval of T_(duration). As the smallest variation that can be determined is a difference of count, this would mean that the smallest change from this base line calibration value that can only be established ⅙th calibration value. This can be made more accurate by designing the pump system to have a high number of clocks during the detection phase by having a lower value of C (the pump coupling capacitance), V (the clock amplitude), or both. For example, designing the pump system to have both C and V reduced by half would increase the number of clocks during the detection phase by a factor of 4, consequently increasing the accuracy by which the load current can be determined. This is a design choice balancing a number of factors, but generally having the system designed so that a higher number of clocks, say 10 or more, during the detection phase can improve the accuracy of the techniques of this section.

Sigma Delta Over-Sampling Charge Pump Analog-to-Digital Converter

This section considers techniques to further improve the accuracy and increase resolution of the of the oversampling charge pump based analog to digital converter (ADC) described in the preceding section. This is done by using the sigma delta concept to noise shape the quantization noise and push this noise into a high frequency range that is out of the signal's bandwidth, allowing higher accuracy and higher resolution. This feature allows for on-chip detection without operation interruption, and allows the detection on the fly (within operation) while still achieving the target accuracy of detection.

For comparison, FIG. 35 looks at the elements of FIG. 33 in some detail. Pump 3001 provides a current I_(output) at a voltage V_(REG) to a load I_(load) 3021. The regulation circulation includes the comparator 3011 that has a first input connected to receive feedback from a node MON between the resistors R1 3013 and R2 3015 and a second input connected to receive a reference voltage V_(REF). From these inputs, the comparator generates the output FLAG, which serves as a first input to the AND gate 3017, that also has an oscillator signal OSC as an input. The output of the AND gate 3017 then is used for the clock signal PUMPCLK of the pump 3001. The ADC Pulse Count block 3007 then is used to determine the current as described in the preceding section.

FIG. 36 shows a first exemplary embodiment that can help to illustrate these various aspects. In FIG. 36 the corresponding elements are numbered as in FIG. 35. In addition to the elements also in FIG. 35, FIG. 36 also includes the elements for implementing sigma delta noise shaping on the output of the pump system. This includes a voltage to current converter Gm1 3031, a capacitor C_(integrate) 3033, and a voltage to current converter Gm2 3035 to form subtraction (delta) between consecutive pulses from the pump, and integration all the difference over C_(integrate) (sigma), and use current summing feedback of I_(FB) at Σ3037 to push the low frequency noise in band to higher frequency range out of signal band width of interest. Here Gm1 3031 and Gm2 3035 are connected in series between the MON node at the input of the comparator 3011 and the adder Σ3037, with the capacitor C_(integrate) 3033 connected between ground and a node between the voltage to current converters.

The sigma delta noise shaping is done by forming the difference (the “delta”) by using Gm1 3031 to convert the sampled voltage into current and using the current's ramp up and ramp down to find the difference between two consecutive cycles. The difference of current is integrated (the “sigma”) over the capacitor C_(integrate) 3033 to obtain low frequency noise voltage. Negative feedback is then used to remove the low frequency error, using Gm2 3035 to convert this low frequency noise voltage back to the current domain (I_(FB)) and summing this with the signal at Σ3037 to remove or reduce the low frequency noise density in the frequency domain, and push the noise power being reduced in low frequency range into the high frequency range, effectively shaping the noise. This is illustrated in FIG. 37.

FIG. 37 shows how the feedback current is summed with signal to remove the low frequency noise away from the consecutive samplings. At top is shown the voltage V_(mon), at the MON node, that is the input to both the comparator 3011 and Gm1 3031, the current I_(Gm1) from Gm1 3031, the voltage VF at the node where C_(integrate) 3033 is connected between the voltage to current converters, and current I_(Gm2) from Gm2 3035. Over time, due to noise shaping, the low frequency noise is reduced. This pushes the quantization noise to high frequencies, where it can be discarded since it is out of signal bandwidth of interest, as discussed further below.

In FIG. 36 the input to the noise shaping section was connected to the feedback loop of the regulation circuit at the input node MON of the comparator 3011, but other arrangements can be used. FIG. 38 shows a variation where the input of Gm1 3031 is now connected between the output of the comparator 3011 and input of the AND gate 3017 to receive the digital pulse FLAG value. As before, Gm1 3031, C_(integrate) 3033, Gm2 3035 to form the subtraction (delta) between two consecutive pulses, perform an integration of all the difference over C_(integrate) 3033 (Sigma), and use current summing feedback to push the low frequency noise in band to a higher frequency range out of signal bandwidth of interest.

FIG. 39 looks at the effect of over-sampling on the regulated voltage level V_(REG) at the output of the pump as supplied to the load. The top waveform shows the fluctuations of V_(REG) about the desired regulation level over time and the bottom shows the corresponding times (t1, t2, . . . ) when the pump clock PUMPCLK is active. The PUMPCLK period variation (such as dT1=t2−t1, dT2=t3−t2, . . . ) is due to noise (such as thermal, 1/f, or quantization noise). Without over-sampling, quantization noise due to the comparator 3011 can have a significant contribution value to the low frequency spectrum, dominating over thermal, 1/f, or other noise within the signal band width. Over-sampling can lower the quantization noise floor over the entire frequency range, which effectively lowers the total noise within the signal bandwidth after decimation filtering. For example, if the ADC pulse counter 3007 (FIG. 36 or 38) achieves (without over-sampling) 4 bits of resolution, 16× over-sampling can provide an additional 4 bits of information, leading to an effective number of bits (ENOB) of 5, or an extra 1 bit of accuracy. This can be further improved through sigma delta over-sapling by the use noise shaping as described in the preceding paragraphs: with 16× sigma delta over-sampling, an extra 3 bits of accuracy, or ENOB-8, can be achieved. This allows for the amount of current being drawn by the pump to be determined with even higher resolution and accuracy.

Consequently, for any of the embodiments of this section, the use of sigma delta over-sampling for further increase the accuracy of charge pump based current detection. Through the subtraction and integration elements illustrated with respect to FIGS. 36 and 38, the ENOB (effective number of bits) can be significantly increased due to noise shaping.

CONCLUSION

Although the various aspects of the present invention have been described with respect to certain embodiments, it is understood that the invention is entitled to protection within the full scope of the appended claims. 

It is claimed:
 1. A circuit connectable to a load to determine the amount of current drawn by the load, comprising: a charge pump system, including: a charge pump connected to receive a clock signal and having an output connectable to provide an output current to the load; and regulation circuitry connected to receive feedback from the output of the charge pump and regulate the charge pump by varying the clock signal based upon the feedback; an output noise shaping circuit connected between the feedback path of the regulation circuitry and the output of the charge pump to provide current feedback to the output of the charge pump to remove low frequency noise; and an analog to digital converter detection circuit connectable to the charge pump system to determine the number of cycles of the clock signal supplied to the charge pump during an interval of a first duration while the charge pump is under regulation and, from the determined number of cycles, determine an amount of current being drawn by the load.
 2. The circuit of claim 1, wherein the output shaping circuit comprises: a first voltage to current converter having an input connected to the feedback path of the regulation circuitry; a second voltage to current converter having an input connected to the output of the first voltage to current converter and an output connected to provide current feedback to the output of the charge pump; and a capacitor connected between ground and a node through with the first and second voltage to current converters are connected.
 3. The circuit of claim 1, wherein the regulation circuitry includes: a comparator having a first input connected to receive a reference voltage and a second input connected to receive the feedback from the output of the charge pump, where the clock signal is varied based upon the output of the comparator.
 4. The circuit of claim 3, wherein the noise shaping circuit is connected to the feedback path of the regulation circuitry at the second input of the comparator.
 5. The circuit of claim 3, wherein the noise shaping circuit is connected to the feedback path of the regulation circuitry at the output of the comparator.
 6. The circuit of claim 3, wherein the second input of the comparator is connected to a node of a voltage divider connected between the output of the charge pump and ground.
 7. The circuit of claim 3, wherein the output of the comparator is connected as a first input to an AND gate, and a second input of the AND gate is connected to receive an oscillator signal, where the clock signal is based on the output of the AND gate.
 8. The circuit of claim 7, wherein the detection circuit has an input connected to the output of the AND gate.
 9. The circuit of claim 8, wherein the detection circuit includes a pulse counter.
 10. The circuit of claim 1, wherein the circuit connectable to a load to determine the amount of current drawn by the load is formed on an integrated circuit including load, the integrated circuit further including: connection circuitry, whereby the output of the charge pump is connectable to drive the load.
 11. The circuit of claim 10, wherein said determination of the amount of current being drawn includes converting the number cycles relative to the reference value to a current value.
 12. The circuit of claim 10, wherein based upon the amount of current being drawn by the circuit component, the current detection circuitry determines whether the circuit component is leaking.
 13. The circuit of claim 10, wherein the current detection circuitry determines the amount of current being drawn by the component by comparing the determined number of cycles to a reference value and, based upon the comparison, determining the amount of current being drawn by the circuit component.
 14. The circuit of claim 10, wherein the charge pump is a negative voltage charge pump.
 15. The circuit of claim 10, wherein the first duration is of a fixed value.
 16. The circuit of claim 10, wherein the integrated circuit includes a non-volatile memory array.
 17. The circuit of claim 16, wherein said circuit component is a word line of the memory array.
 18. The circuit of claim 10, wherein the integrated circuit includes a DRAM memory array.
 19. The circuit of claim 18, wherein said current is an amount of background leakage of the DRAM memory array. 